
==== Front
iScience
iScience
iScience
2589-0042
Elsevier

S2589-0042(24)01974-6
10.1016/j.isci.2024.110749
110749
Article
High isolation circularly polarized MIMO antenna based on metasurfaces
Wu Ting wutingzdh@xaut.edu.cn
123∗
Yan Lei 1
1 Xi’an Key Laboratory of Wireless Optical Communication and Network Research School of Automation and Information Engineering, Xi’an University of Technology, Xi’an, Shaanxi Province 710048, China
2 Electronic Science and Technology Postdoctoral Research Center, School of Automation and Information Engineering, Xi’an University of Technology, Xi’an, Shaanxi Province 710048, China
∗ Corresponding author wutingzdh@xaut.edu.cn
3 Lead contact

21 8 2024
20 9 2024
21 8 2024
27 9 11074921 6 2024
5 8 2024
13 8 2024
© 2024 The Author(s)
2024
https://creativecommons.org/licenses/by-nc-nd/4.0/ This is an open access article under the CC BY-NC-ND license (http://creativecommons.org/licenses/by-nc-nd/4.0/).
Summary

In this article, a metasurface (MS) for decoupling circularly polarized (CP) antenna arrays is presented. The MS consists of periodic Jerusalem cross slot on one side of the substrate. MS can suppress space wave coupling by changing the coupling path. The distance between the centers of the antenna units is 30 mm (0.35λ0). A 2 × 2 CP microstrip array coupled antenna is simulated and fabricated. The experimental results show that the mutual coupling between the E-plane and H-plane is reduced by 24 dB and 16 dB at the center frequency. It is worth mentioning that the antenna gain has been improved by 1.5 dBi.

Graphical abstract

Highlights

• Simultaneous decoupling of E-plane and H-plane of circularly polarized arrays

• The decoupled structure did not destroy the impedance of the original arrays

• Arrays are periodically arranged to facilitate further expansion

• The gain of the antenna array is improved by loading the decoupling structure

Applied sciences; Photonics; Devices

Subject areas

Applied sciences
Photonics
Devices
Published: August 21, 2024
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pmcIntroduction

With the rapid development of the electronic information industry in the 21st century, wireless communication has been particularly prominent in this field. Nowadays, the Internet of Things is increasingly integrated with communications technology and has become one of the core technologies. Multiple input and multiple output (MIMO) antenna technology is being applied more and more widely. However, with the increase in the functions that the antenna needs to achieve, the design of the antenna is becoming more and more demanding. This requires increasing the number of antennas to accomplish these functions. However, as the number of antennas increases and the spacing decreases, it will cause coupling between MIMO antennas. This will cause the impedance matching performance of the array element to worsen and the main beam of the unit pattern to narrow.1 The best solution is to increase the distance between the units, but this contradicts miniaturization. Therefore, it is necessary to reduce the coupling between the antennas to improve the antenna’s performance. How to increase the isolation of the antenna has become a research focus of MIMO antennas.

The coupling between the two antennas mainly comes from three parts– the space wave coupling between the antenna elements, the surface wave coupling produced by the shared substrate between the antenna elements, and the surface electric current coupling between antenna units due to shared ground. Therefore, decoupling research is conducted from both spatial and ground directions. Currently, we already know quite a few decoupling methods.

Defected ground structure (DGS),2,3,4 neutralization-line(NL),5,6,7 electromagnetic band gap (EBG),8,9,10 decoupling network (DN),11,12,13 metasurface antenna array decoupling (MAAD),14,15,16 etc. A DGS was proposed in Qian et al.2 using common-mode (CM) differential-mode (DM) theory with the help of machine learning to suppress both E-plane and H-plane coupling. In Wu and Wang,5 a new coupling path is created by adding a neutralizing line between the two antenna units, thus reducing the mutual coupling between the antennas, which ultimately results in an isolation of 19.2 dB, 22.26 dB, and 17.9 dB between orthogonally and parallelly placed antenna elements, respectively. A novel mushroom-shaped EBG structure was proposed in Tan et al.8 By opening another gap on the surface of the original EBG structure and utilizing its two resonant modes, the coupling of the antenna at 3.48 GHz and 4.88 GHz is reduced by 26 dB and 44 dB respectively. It is proved that the structure has the characteristics of surface wave suppression. An efficient T-decoupling network is proposed in Zou et al.11 to decouple closely spaced dual-element E-plane antenna arrays, which are subsequently extended to multivariate linear transmission lines with good decoupling results. MS was used for the first time in Wang et al.14 for decoupling between antennas, where decoupling is achieved by placing a split ring resonator on top of the antenna array, resulting in an increase in isolation between the antennas from 8 dB to 27 dB, as well as an improvement in gain.

In this article, a loaded MS high isolation CP MIMO antenna is proposed. First, the antenna is powered by a 90-degree phase-shifted Wilkinson power divider in the bottom layer to complete the left-hand circular polarization (LHCP). To solve the problem of poor isolation between the antenna units, the isolation between the antenna units is effectively improved by loading the MS, which consists of a metal-etched periodic Jerusalem slit on one side of the dielectric plate and is located above the CP array. At the center frequency, the E-plane and H-plane mutual coupling is reduced by 24 dB and 16 dB respectively, At the same time, good CP performance was obtained. The measured results agree well with the simulation results.

The characteristics of this structure:(1) The proposed MS can simultaneously reduce coupling in both E-plane and H-plane.

(2) The MS does not disrupt the original antenna’s impedance; no need for further adjustments.

(3) The dimensions of the MS and the antenna are designed periodically, facilitating their use in larger arrays.

Design of antenna structure

Metasurface decoupling principle and unit design

Figure 1 shows the schematic diagram of MS decoupling. when the MS is not loaded, the electromagnetic wave from the antenna is along the positive direction of the X axis, and the propagation coefficient K of the electromagnetic wave in free space can be expressed as:(Equation 1) k=ωμ·ε

Figure 1 Schematic of MS decoupling

Loading MS with different dielectric constants and permeabilities on the upper surface of the array antenna units will reduce the space wave coupling between the antenna units. From Equation 1 the propagation coefficient of the electromagnetic wave can be expressed as 11 when the MS is loaded:(Equation 2) k=ω·εxε0·μxμ0=jk0|μx|·|εx|

The formula can be further evolved:(Equation 3) E(x,t)=E0ejkxejωt=E0ej·(jk0·|μx||εx|)·x·ejkt=E0e−k0|μx||εx|·x·ejωt

Equation 3 can be introduced when the propagation of electromagnetic waves along the X axis direction gradually decreases; at this time the electromagnetic waves mainly propagate along the z axis, so it is only necessary to array the antenna above the loading of negative permittivity (εx < 0) and positive permeability (μx > 0) MS can be.

To find the optimal parameters of εxμx, using ideal homogeneous metamaterials (MTM), the coupling situation of the antenna was observed by simulating MTM superstrate covering over the antenna array with HFSS. According to (Figure 2A), the antenna is printed on an FR4 substrate with a thickness of 1.6 mm. The side length of the antenna is 19.4 mm, and the distance between the antenna units is 30 mm (0.35λ0). The antenna is symmetrically placed along the XOZ. The volume of the MTM is = 75 mm × 45 mm × 30 mm. An air gap of Hs = 2 mm is inserted between the dielectric plate and the MTM superstrate. Change the dielectric constant and magnetic permeability of the MTM superstrate in CST. As shown in (Figure 2B) of the simulation results, when εxμx = 3.55, the transmission coefficient S12 of the antenna is minimal at 3.5 GHz, meaning the coupling between the array antenna units is at its minimum.Figure 2 MTM superstrate

(A) Load the array antenna of the MTM superstrate.

(B) Ideal model transmission coefficient.

Design MS that satisfies εxμx, as shown in (Figure 3A), the MS unit is printed on Rogers4350B with a thickness of 1.524 mm. The unit is composed of a Jerusalem cross gap etched in the metal layer, with the side length of the unit being P = 15 mm, the length of the main gap being Ls1 = 14 mm, the length of the extension gap being Ls2 = 10 mm, and the width of the gap being W1 = 1 mm. Set the radiation boundaries on all four sides of the MS unit to two pairs of master-slave boundary conditions using HFSS, and set a pair of Floquet ports on the XOY plane, as shown in (Figure 3B) Through simulation, the S-parameters of the periodic structure can be obtained as shown in (Figure 3C). Infer the εx and μx of the MS unit through S-parameters. (Figure 3D), it can be seen that at 3.5 GHz, εx is negative and μx is positive for the MS unit.Figure 3 MS unit

(A) Top view of the MS.

(B) Master-slave boundary and Floquet port settings.

(C) S-parameters of the periodic structure.

(D) εx and μx.

The process of S-parameter inversion εx and μx is as follows:(Equation 4) S21=S12=1[sin(nkd)−i2(Z+1Z)cos(nkd)]eikd

(Equation 5) S11=S22=i2(1Z−Z)sin(nkd)

Where n is the refractive index. d represents the thickness of the material. Z represents impedance. It can be deduced from (Equation 4) and (Equation 5) that n and Z can be represented as(Equation 6) n=1kdcos−1[12S21(1−S112+S212)+2πm]

(Equation 7) Z=(1+S112)−S212(1−S112)−S212

So, εx and μx can be represented as:(Equation 8) μx=nZ

(Equation 9) εx=n/Z

1 × 2 H-Plane coupled antennas

Figure 4 is the H-plane coupled antenna array with two elements. The top layer is a Rogers 4350B (εr = 3.66, tanδ = 0.004) substrate with a thickness of 1.524 mm. The designed MS cells are printed on the substrate in a 3 × 5 arrangement. A slight modification of Ls1 is required here. The middle layer is made of FR4 (εr = 4.4, tanδ = 0.02) with a thickness of 1.6 mm. A square patch with side length Wp is printed on its upper surface and the center distance of the patch is 30 mm. On the lower surface, there are four circular gaps of R1. The bottom layer is a Rogers 4350B with a thickness of 0.508 mm. A 90-degree phase-shifted Wilkinson power divider is printed on the undersurface of the substrate. The microstrip line is terminated with a 50 Ω SMA. Table 1 contains the specific parameters.Figure 4 Structures of H-plane arrays

(A) 3D view of the antenna.

(B) H-plane coupling array top view of the middle layer.

(C) Wilkinson power divider network layer.

Table 1 Dimensions of H-surface coupled patch and MS (unit: mm)

Lg	Wg	Wp	Xf	D	Ls1	Ls2	W1	R0	H1	
75	40	19.4	3.8	30	14.5	11.6	1	0.45	1.524	
H2	H3	H4	R1	R2	W2	W3	L1	L2	L3	
6	1.6	0.508	1.05	1	1.06	0.56	16.8	5.94	2	
L4	L5	L6	L7	L8						
4.38	4.36	1.94	6.2	6.53						

The S-parameters, gain, and axial ratios (AR) of the H-plane array with unloaded MS and loaded MS are shown in Figure 5 The loading of the MS changes the propagation mode of the electromagnetic wave, which makes the space-wave coupling in the H-plane between antenna units greatly reduced. The simulation results show that S12 is reduced from −17.3 dB to −39.5 dB at the center operating frequency and the mutual coupling is reduced by more than 20 dB. In addition, the MS also plays the role of guiding, so the gain is increased from 3.1 dBi to 4.5 dBi. The MS is still some distance away from the antenna layer, which has little effect on the antenna’s polarization mode, and the AR is less than 3 dB over the whole band, which has good CP characteristics.Figure 5 Comparison of H-plane parameters without MS and with MS

(A) S- parameters.

(B) Gain.

(C) AR.

1 × 2 E-Plane coupled antennas

The E-plane coupled array is shown in Figure 6 using the same decoupling structure. For the convenience of further work, the distance between the centers of the patches remains D, and the same number of units is also set for the top layer of the MS.Figure 6 Structures of E-plane arrays

(A) 3D view of the antenna.

(B) E-plane coupling array top view of the middle layer.

The E-plane coupled array simulation parameters are shown in Figure 7. It can be observed that the loaded MS still acts as a spatial filter in the E-plane. The S12 decreases from −18 dB to −26 dB at the center frequency, and the MS also has a directional effect that increases the gain from 3.2 dBi to 4.6 dBi. The axial ratio is less than 3 dB in the frequency band, and it still maintains a good circular polarization characteristic.Figure 7 Comparison of E-plane parameters without MS and with MS

(A) S- parameters.

(B) Gain.

(C) AR.

2 × 2 coupled antennas

Combine the antennas proposed in Sections 2.2 and 2.3 to form a new 2 × 2 antenna arrays. The structure of the antenna is shown in Figure 8. For convenience of measurement, the ground of the array antenna is short-circuited to the upper layer of the middle layer substrate through a metal hole with a radius of R4. The red part is 100Ω resistance, the yellow part is the feed network, and the blue part represents where the SMA is welded, secured all around with a radius R3 of Teflon. To achieve better isolation at the center frequency, it is necessary to fine-tune the gaps in the MS, finally determining Ls2 = 11.3 mm.Figure 8 Structures of 2 × 2 arrays

(A) 3D view of the antenna.

(B) 2 × 2 coupling array top view of the middle layer. Parameters:R3 = 0.4 mm, R4 = 1 mm, X1 = 9.86 mm, Y1 = 5 mm.

Since the 2 × 2 arrays antennas are symmetrically placed, antenna 1 is chosen here as a representative for analysis. The excitation is added to port 1 and the other ports are connected to 50 Ω loads. From Figure 9, it can be seen that S12 decreases from −18 dB to −42 dB, S13 decreases from −17 dB to −33 dB, and S14 decreases from −23 dB to −34 dB after loading the MS at the center frequency. The gain of antenna 1 is increased from 2.5 dBi to 3.9 dBi. For the gain of the entire array, the antenna increases across the entire frequency band after loading MS. The antenna has a good CP characteristic with an AR of less than 3 dB in the operating frequency band.Figure 9 Comparison of 2 × 2 parameters without MS and with MS

(A) S-parameters.

(B) Antenna 1 gain.

(C) The gain of the whole array.

(D) AR.

Parametric studies

To get the final results, each parameter was simulated and analyzed to determine the current value. After scanning all parameters, it was found that the air layer height H2 and gap LS2 on the MS have the greatest impact on the isolation.

Figure 10 shows the simulation results of different H2 values. It can be observed that the change of H2 has little effect on S11 but significantly impacts S12, S13, and S14. From the changes in S12 and S13, it can be concluded that H2 affects the maximum isolation of the center frequency, while the decoupling center frequency does not change significantly. From the changes in S14, it can be concluded that H2 affects the decoupling center frequency. Choose appropriate parameters to minimize the coupling between the E-plane and H-plane, finally determining H2 = 6 mm.Figure 10 Simulate the S-parameters of different H2 values

(A) S11.

(B) S12.

(C) S13.

(D) S14.

As shown in Figure 11, unlike H2, increasing the value of the Jerusalem gap LS2 will shift the decoupling center frequency toward a lower frequency. This is because the length of the gap can change the resonant frequency of the MS. Similarly, choosing the appropriate LS2 ensures maximum isolation of the E-plane and H-plane.Figure 11 Simulate the S-parameters of different Ls2 values

(A) S11.

(B) S12.

(C) S13.

(D) S14.

Results and discussion

Make and measure the antenna as mentioned above as shown in Figure 12. The dimensions of the antenna are 75 mm × 75 mm × 9.632 mm (0.87λ0 × 0.87 λ0 × 0.11 λ0). (Figure 13A) shows the simulated and measured S-parameters of the array antenna. The simulated and measured results are similar. The differences arise due to the manufacturing and measurement tolerances of the antenna. At the same time, the results of measurements gain and AR match well with the simulation, which further demonstrates the effectiveness of our design.Figure 12 Photographs of the antenna with MS and without MS

(A) Without MS.

(B) With MS.

Figure 13 Comparison of simulation and measurement parameters

(A) S- parameters.

(B) Gain.

(C) AR.

Figure 14 illustrates the normalized radiation direction maps for LHCP and right-hand circular polarization (RHCP) with and without MS loading. The results analyzed after comparison are as follows.Figure 14 LHCP and RHCP radiation patterns

(A) Simulated E-plane without MS and with MS.

(B) Simulated H-plane without MS and with MS.

(C) Simulated and measured radiation patterns of E-plane.

(D) Simulated and measured radiation patterns of H-plane.

For the LHCP radiation patterns

The normalized backward gains with and without MS are −20.5 dBi and −22.4 dBi respectively. In the H-plane, the maximum gain of the antenna is not in the +Z direction because the antenna units are too close to each other, so it results in the maximum gain of the antenna in the +Z direction and the maximum gain is θ = 30°; in short, loading of the MS does not have a great effect on the LHCP radiation direction graph.

For the RHCP radiation patterns

The normalized forward gains with and without MS are −24.8 dBi and −18 dBi respectively, and the reverse gains are −20 dBi and −27 dBi respectively. From the results, it can be seen that the forward cross-polarized gain is reduced by 6 dBi and the reverse cross-polarized gain is increased by 7 dBi after adding the MS.

The far-field radiation direction plot shows that the effect on the main polarisation is not very large and does not deteriorate the front-to-back ratio of the antenna after loading the MS. The slightly larger effect on the cross-polarization is due to the improved AR of the antenna at the center frequency after MS loading. The measured results are very close to the simulation results. Table 2 compares the designed antenna with the current decoupling methods reported in the literature. Compared with other works of literature, the MS designed in this article can reduce both E-plane and H-plane coupling without destroying the performance of the original antenna and has good decoupling performance.Table 2 Comparison results of studies of others and this study

Freq. (GHz)	Isolation value(dB)	Gap	Method	Achievements	References	
2.45	40	0.37λ0	DGS	Reduce E− and H-plane couplings	Qian et al2	
3.5/4.9	26/44	40mm	EBG	Dual frequency band	Tan et al8	
3.5	40	0.5λ0	NL	Reduce E− and H-plane couplings	Qi et al12	
1.268	25	0.4λ0	Wall	circularly polarized antenna	Zhang et al17	
3.5	40	0.4λ0	MS	Reduce H-plane couplings + Gain improve	Guo et al18	
3.5	42	0.35λ0	MS	Reduce E− and H-plane couplings + Gain improve + circularly polarized antenna	This work	

Conclusion

In this article, an MS for decoupling CP antenna arrays is designed. The MS can be used as a spatial filter to suppress the coupling between E-plane and H-plane cells. By loading MS on top of 1 × 2 H plane arrays, 1 × 2 E plane arrays, and 2 × 2 arrays, verify that MS has a good decoupling effect. The final antenna was machined and measured. Simulations and measurements show that the isolation between the antenna units is greatly improved. In addition, the antenna still has good CP characteristics and the gain is improved. This antenna has excellent application scenarios in the future communication field.

Limitations of the study

For the proposed high isolation CP MIMO antenna array, the gain is not as high as desired. The main reason is that the distance is too close and the directional map is not well improved. But the whole array is set up periodically and the total number of arrays can be increased to enhance the gain. In addition, the size of the entire antenna is tiny and the design is susceptible to the manufacturing tolerances, which requires a high level of manufacturing accuracy to reproduce the same results.

Resource availability

Lead contact

Further information and resources related to this study will be fulfilled by the lead contact Ting Wu (wutingzdh@xaut.edu.cn) upon reasonable request.

Materials availability

This paper did not generate new unique reagents.

Data and code availability

Data reported in this paper will be shared by the lead contact upon request.

This paper does not report the original code.

Any additional information required to reanalyze the data reported in this paper is available from the lead contact upon request.

Acknowledgments

This work was supported in part by 10.13039/501100001809 National Natural Science Foundation of China under Grant 62202372 ; Young Talent fund of University Association for Science and Technology in Shaanxi, China under Grant 20200111 ; Key Research and Development Plan of Shaanxi Province (General project) under Grant 2023-YBGY-038 ; Shaanxi Key Laboratory of Space Extreme Detection Fund under Grant SED20221203 ; Program for Talent of Colleges and Universities Service Enterprise of Xi’an under Grant 23GXFW006 . We appreciate the editor’s and reviewers’ valuable comments and suggestions for our manuscript. All data generated or analyzed during this study are included in this published article.

Author contributions

Methodology, T.W.; Investigation, Writing Review and Editing, Y.L.

Declaration of interests

The authors declare no competing financial interests.

STAR★Methods

Key resources table

REAGENT or RESOURCE	SOURCE	IDENTIFIER	
Software and algorithms	
	
Ansys Electronics Desktop R2023b	High Frequency Structure Simulator	https://www.ansys.com/products/electronics/ansys-hfss	
CST Studio Suite 2023	Computer Simulation Technology	https://www.3ds.com/products/simulia/cst-studio-suite	
	
Other	
	
Vector Network Analyzer	Keysight E5071C	https://www.keysight.com/zz/en/products/network-analyzers.html	

Method details

Numerical analysis

All numerical analysis for this work was performed in Ansys Electronics Desktop R2023b and CST Studio Suite 2023.In this paper, a loaded MS high isolation CP MIMO antenna is proposed. Firstly, the antenna is powered by a 90 -degree phase-shifted Wilkinson power divider in the bottom layer to complete the left-hand circular polarisation (LHCP). To solve the problem of poor isolation between the antenna units, the isolation between the antenna units is effectively improved by loading the MS, which consists of a metal-etched periodic Jerusalem slit on one side of the dielectric plate and is located above the CP array. At the center frequency, the E-plane and H-plane mutual coupling is reduced by 24 dB and 16 dB respectively, At the same time, good CP performance was obtained. The measured results agree well with the simulation results.

Sample fabrication

Make and measure the antenna as mentioned above as shown in Figure 12. The dimensions of the antenna are 75 mm × 75 mm × 9.632 mm (0.87λ0 × 0.87 λ0 × 0.11 λ0). For convenience of measurement, the ground of the array antenna is short-circuited to the upper layer of the middle layer substrate through a metal hole with a radius of R4.
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